Orthogonal frequency division multiplexing synchronization

ABSTRACT

Embodiments are directed to first and second OFDM pilot symbols. The first and second pilot symbols may have first and second sets, respectively, of allowed, forbidden, and active carrier frequencies. The second sets of carrier frequencies may be formed by frequency shifting the respective first sets by a predetermined frequency, such as the frequency difference between adjacent carriers. An embodiment is directed to frequency translating part of a first received pilot symbol by one carrier interval in a first direction, frequency translating part of a second received pilot symbol by one carrier interval in a second direction that is opposite from the first direction, and forming a correlation by multiplying the frequency translated parts of the first and second pilot symbols by complex conjugates of parts of the pilot symbols upon which frequency translation has not been performed, and summing the multiplication results.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a divisional of U.S. patent application Ser. No.11/934,462, filed Nov. 2, 2007, and titled “Orthogonal FrequencyDivision Multiplexing Synchronization,” the contents of which are herebyincorporated by reference in their entirety.

FIELD OF THE INVENTION

Embodiments relate generally to communications networks. Morespecifically, embodiments relate to orthogonal frequency divisionmultiplexing synchronization.

BACKGROUND OF THE INVENTION

Digital broadband broadcast networks enable end users to receive digitalcontent including video, audio, data, and so forth. Using a mobileterminal, a user may receive digital content over a wireless digitalbroadcast network. Digital content can be transmitted in a cell within anetwork. A cell may represent a geographical area that may be covered bya transmitter in a communication network. A network may have multiplecells and cells may be adjacent to other cells.

A receiver device, such as a mobile terminal, may receive a program orservice in a data or transport stream. The transport stream carriesindividual elements of the program or service such as the audio, videoand data components of a program or service. Typically, the receiverdevice locates the different components of a particular program orservice in a data stream through Program Specific Information (PSI) orService Information (SI) embedded in the data stream. However, PSI or SIsignaling may be insufficient in some wireless communications systems,such as Digital Video Broadcasting-Handheld (DVB-H) systems. Use of PSIor SI signaling in such systems may result in a sub-optimal end userexperience as the PSI and SI tables carrying in PSI and SI informationmay have long repetition periods. In addition, PSI or SI signalingrequires a large amount of bandwidth which is costly and also decreasesefficiency of the system.

BRIEF SUMMARY OF THE INVENTION

The following presents a simplified summary in order to provide a basicunderstanding of some aspects of the invention. The summary is not anextensive overview of the invention. It is neither intended to identifykey or critical elements of the invention nor to delineate the scope ofthe invention. The following summary merely presents some concepts ofthe invention in a simplified form as a prelude to the more detaileddescription below.

Embodiments are directed to first and second OFDM pilot symbols. Thefirst and second pilot symbols may have first and second sets,respectively, of allowed, forbidden, and active carrier frequencies. Thesecond sets of carrier frequencies may be formed by frequency shiftingthe respective first sets by a predetermined frequency, such as thefrequency difference between adjacent carriers. An embodiment isdirected to frequency translating part of a first received pilot symbolby one carrier interval in a first direction, frequency translating partof a second received pilot symbol by one carrier interval in a seconddirection that is opposite from the first direction, and forming acorrelation by multiplying the frequency translated parts of the firstand second pilot symbols by complex conjugates of parts of the pilotsymbols upon which frequency translation has not been performed, andsumming the multiplication results.

BRIEF DESCRIPTION OF THE DRAWINGS

A more complete understanding of the present invention and theadvantages thereof may be acquired by referring to the followingdescription in consideration of the accompanying drawings, in which likereference numbers indicate like features, and wherein:

FIG. 1 illustrates a suitable digital broadband broadcast system inwhich one or more illustrative embodiments of the invention may beimplemented.

FIG. 2 illustrates an example of a mobile device in accordance with anaspect of the present invention.

FIG. 3 illustrates an example of cells schematically, each of which maybe covered by a different transmitter in accordance with an aspect ofthe present invention.

FIG. 4 shows a frame and superframe of symbols, synchronization symbolsused for channel searches and service discovery, and data in accordancewith an aspect of the invention.

FIG. 5 shows how a signal center frequency may coincide with, or beoffset relative to, a channel center frequency.

FIG. 6 is a flow chart showing steps performed by a receiver inaccordance with at least one embodiment.

FIG. 7 shows an example of the size of a pilot signal bandwidth relativeto a signal bandwidth and a channel raster bandwidth in accordance withan aspect of the invention.

FIG. 8 illustrates sparse pilot spacing of a pilot sequence for a pilotsymbol in accordance with an aspect of the invention.

FIG. 9 is a flowchart showing steps performed by a receiver forperforming correlation in the frequency domain to detect the coarseoffset being used.

FIG. 10 is a flow chart that shows steps in accordance with anembodiment for performing a service discovery correlation in the timedomain.

FIG. 11 shows an example of a pilot/signaling symbol sequence inaccordance with an aspect of the invention.

FIG. 12 is a flowchart showing steps of a method performed by atransmitter in accordance with at least one aspect of the invention.

FIGS. 13 and 14 depict relationships between P1, P2, and DATA symbols inaccordance with an aspect of the invention.

FIG. 15 shows an exemplary frame and slot structure including OFDMsymbols and cells in accordance with an aspect of the invention.

FIG. 16 illustrates coherence bandwidth and differential modulationwithin one pilot symbol.

FIG. 17 depicts differential modulation between two P1 symbols inaccordance with an aspect of the invention.

FIG. 18 shows two 1 k symbols with 1/1 guard interval and differentialmodulation between the symbols in accordance with an embodiment.

FIG. 19 shows calculation of sums of received energy from one or morepilot symbols in accordance with one or more embodiments.

FIG. 20 shows a transmitter in accordance with one or more embodiments.

FIG. 21 shows a receiver in accordance with one or more embodiments.

FIG. 22 is a flow diagram that shows steps that may be performed by areceiver in accordance with one or more embodiments.

FIG. 23 is a graph of auto/cross-correlations between pilot sequencesand their frequency offset versions in accordance with one or moreembodiments.

FIG. 24 is a zoomed in version of the graph of FIG. 23 showing the lowcross-correlation range of frequency offsets.

FIG. 25 is a graph that shows the envelope amplitude of a first pilotsymbol signal in accordance with at least one embodiment.

FIG. 26 is a zoomed in version of the graph of FIG. 25.

FIG. 27 shows an example of a 2 k symbol (P1) symbol, in accordance withan embodiment.

FIG. 28 shows a synchronization symbol P1 having two consecutive OFDMsymbols (P1a, and P1b) that have the same FFT size in accordance with anembodiment.

FIG. 29 shows an example of P1 in which pulses P1a and P1b have eachbeen subdivided into two parts in accordance with an embodiment.

FIG. 30 is a schematic diagram of a correlator portion of a receiver inaccordance with an embodiment.

FIG. 31 is a schematic diagram of a correlator portion of a receiver inaccordance with an embodiment.

FIG. 32 shows steps of a detection sequence in accordance with anembodiment.

DETAILED DESCRIPTION OF THE INVENTION

In the following description of the various embodiments, reference ismade to the accompanying drawings, which form a part hereof, and inwhich is shown by way of illustration various embodiments in which theinvention may be practiced. It is to be understood that otherembodiments may be utilized and structural and functional modificationsmay be made without departing from the scope and spirit of the presentinvention.

Embodiments are directed to service discovery and channel search indigital broadcast networks. Relatively fast service discovery isdesirable from a user's point of view. Naturally, the first timereceiver device is used, a blind service discovery/channel search isperformed. Also, when a terminal is switched off and moved to adifferent location, a new blind search is also performed. Usually, amobile TV application also runs background channel search from time totime in order to detect possible new services. The blind servicediscovery should only take a couple of seconds so as not to irritate theend user and to enable frequent re-scans.

The challenges related to conventional digital video broadcast servicediscovery include the following. The DVB-H standard offers a lot offlexibility for the signal bandwidths, FFT sizes, Guard Intervals, Innermodulations and the like. Operators may use offsets for the DVB-Hsignal, i.e., the signal is not at the nominal center frequency of achannel, but is offset a certain amount. Different countries usedifferent channel raster and signal bandwidth. TPS (TransmitterParameter Signaling) is included in the standard to help receiversynchronization and channel search. Unfortunately, the receiver needs toknow several parameters before it can decode TPS information. Signalbandwidth, frequency offset, FFT size, and Guard Interval need to beknown before the TPS can be decoded. Most of the channels in the UHFband do not contain DVB-H service. The non-DVB-H channels are detectedwith a trial-and-error method (trying to achieve lock with all modes),and that consumes a lot of time. The time to detect non-DVB-H servicesactually mainly sets the achievable speed for channel search becauseusually most of the channels are empty or contain non-DVB-H service.

An example calculation for the blind service discovery is as follows:number of channels in UHF 35, (Channels 21-55, 470-750 MHz); number offrequency offsets 7 (− 3/6, − 2/6, −⅙, 0, +⅙, + 2/6, + 3/6 MHz); numberof signal bandwidths 3 (6 MHz, 7 MHz, 8 MHz. 5 MHz is separate case onlyfor USA receivers); number of FFT sizes 3 (2K, 4K, 8K); number of GuardIntervals 4 ( 1/32, 1/16, ⅛ and ¼); and average time to decode TPS forone mode 120 ms (2K 50 ms, 4K 100 ms, 8K 200 ms). The numbers areexemplary.

The resulting time for blind service discovery would be in this example:35*7*3*3*4*120 ms=1058.4 seconds=17.64 minutes.

In accordance with embodiments, various methods may be used to reducehow long it takes to perform channel search/service discovery. The basicidea of the various methods is to introduce a part of a signal (e.g.initialization/synchronization symbol(s)), which has knowncharacteristics and remains the same with different digital videobroadcast operation modes. Therefore, the known part of the signal canbe decoded without having to resort to trial and error methods. Theknown part of signal contains the parameters for the rest of the signal;therefore, the rest of the signal can be decoded without trial and errormethods after the known part is decoded. The known part of the signalcomprises a subset of available subcarriers and their modulation. Thecombination of the predefined subcarriers (subcarrier numbers) and theirmodulation is chosen so that the combination is unique for eachoffset-FFT size pair (or unique for the different FFT-sizes only) andwhich combination may be used for identifying the signal as a desiredsignal for the digital video broadcast. Also, the channels containingdigital video broadcast services can be efficiently detected using theknown part of the signal. If the fixed known part is not found from theexamined signal, then the signal will be considered anon-digital-video-broadcast signal or an empty channel, and the receivercan promptly proceed to a next channel/frequency. In this way, detectingnon-digital-video-broadcast and empty channels becomes relatively fast.

FIG. 1 illustrates a suitable digital broadband broadcast system 102 inwhich one or more illustrative embodiments may be implemented. Systemssuch as the one illustrated here may utilize a digital broadbandbroadcast technology, for example Digital Video Broadcast-Handheld(DVB-H) or next generation DVB-H networks. Examples of other digitalbroadcast standards which digital broadband broadcast system 102 mayutilize include Digital Video Broadcast-Terrestrial (DVB-T), IntegratedServices Digital Broadcasting-Terrestrial (ISDB-T), Advanced TelevisionSystems Committee (ATSC) Data Broadcast Standard, Digital MultimediaBroadcast-Terrestrial (DMB-T), Terrestrial Digital MultimediaBroadcasting (T-DMB), Satellite Digital Multimedia Broadcasting (S-DMB),Forward Link Only (FLO), Digital Audio Broadcasting (DAB), and DigitalRadio Mondiale (DRM). Other digital broadcasting standards andtechniques, now known or later developed, may also be used. Aspects ofthe invention may also be applicable to other multicarrier digitalbroadcast systems such as, for example, T-DAB, T/S-DMB, ISDB-T, andATSC, proprietary systems such as Qualcomm MediaFLO/FLO, andnon-traditional systems such 3GPP MBMS (Multimedia Broadcast/MulticastServices) and 3GPP2 BCMCS (Broadcast/Multicast Service).

Digital content may be created and/or provided by digital contentsources 104 and may include video signals, audio signals, data, and soforth. Digital content sources 104 may provide content to digitalbroadcast transmitter 103 in the form of digital packets, e.g., InternetProtocol (IP) packets. A group of related IP packets sharing a certainunique IP address or other source identifier is sometimes described asan IP stream. Digital broadcast transmitter 103 may receive, process,and forward for transmission multiple digital content data streams frommultiple digital content sources 104. In various embodiments, thedigital content data streams may be IP streams. The processed digitalcontent may then be passed to digital broadcast tower 105 (or otherphysical transmission component) for wireless transmission. Ultimately,mobile terminals or devices 112 may selectively receive and consumedigital content originating from digital content sources 104.

As shown in FIG. 2, mobile device 112 may include processor 128connected to user interface 130, memory 134 and/or other storage, anddisplay 136, which may be used for displaying video content, serviceguide information, and the like to a mobile-device user. Mobile device112 may also include battery 150, speaker 152 and antennas 154. Userinterface 130 may further include a keypad, touch screen, voiceinterface, one or more arrow keys, joy-stick, data glove, mouse, rollerball, touch screen, or the like.

Computer executable instructions and data used by processor 128 andother components within mobile device 112 may be stored in a computerreadable memory 134. The memory may be implemented with any combinationof read only memory modules or random access memory modules, optionallyincluding both volatile and nonvolatile memory. Software 140 may bestored within memory 134 and/or storage to provide instructions toprocessor 128 for enabling mobile device 112 to perform variousfunctions. Alternatively, some or all of mobile device 112 computerexecutable instructions may be embodied in hardware or firmware (notshown).

Mobile device 112 may be configured to receive, decode and processdigital broadband broadcast transmissions that are based, for example,on the Digital Video Broadcast (DVB) standard, such as DVB-H or DVB-T,through a specific DVB receiver 141. The mobile device may also beprovided with other types of receivers for digital broadband broadcasttransmissions. Additionally, receiver device 112 may also be configuredto receive, decode and process transmissions through FM/AM Radioreceiver 142, WLAN transceiver 143, and telecommunications transceiver144. In one aspect of the invention, mobile device 112 may receive radiodata stream (RDS) messages.

In an example of the DVB standard, one DVB 10 Mbit/s transmission mayhave 200, 50 kbit/s audio program channels or 50, 200 kbit/s video (TV)program channels. The mobile device 112 may be configured to receive,decode, and process transmission based on the Digital VideoBroadcast-Handheld (DVB-H) standard or other DVB standards, such asDVB-MHP, DVB-Satellite (DVB-S), or DVB-Terrestrial (DVB-T). Similarly,other digital transmission formats may alternatively be used to delivercontent and information of availability of supplemental services, suchas ATSC (Advanced Television Systems Committee), NTSC (NationalTelevision System Committee), ISDB-T (Integrated Services DigitalBroadcasting-Terrestrial), DAB (Digital Audio Broadcasting), DMB(Digital Multimedia Broadcasting), FLO (Forward Link Only) or DIRECTV.Additionally, the digital transmission may be time sliced, such as inDVB-H technology. Time-slicing may reduce the average power consumptionof a mobile terminal and may enable smooth and seamless handover.Time-slicing entails sending data in bursts using a higher instantaneousbit rate as compared to the bit rate required if the data weretransmitted using a traditional streaming mechanism. In this case, themobile device 112 may have one or more buffer memories for storing thedecoded time sliced transmission before presentation.

In addition, an Electronic Service Guide (ESG) may be used to provideprogram or service related information. Generally, an Electronic ServiceGuide (ESG) enables a terminal to communicate what services areavailable to end users and how the services may be accessed. The ESGincludes independently existing pieces of ESG fragments. Traditionally,ESG fragments include XML and/or binary documents, but more recentlythey have encompassed a vast array of items, such as for example, a SDP(Session Description Protocol) description, textual file, or an image.The ESG fragments describe one or several aspects of currently available(or future) service or broadcast program. Such aspects may include forexample: free text description, schedule, geographical availability,price, purchase method, genre, and supplementary information such aspreview images or clips. Audio, video and other types of data includingthe ESG fragments may be transmitted through a variety of types ofnetworks according to many different protocols. For example, data can betransmitted through a collection of networks usually referred to as the“Internet” using protocols of the Internet protocol suite, such asInternet Protocol (IP) and User Datagram Protocol (UDP). Data is oftentransmitted through the Internet addressed to a single user. It can,however, be addressed to a group of users, commonly known asmulticasting. In the case in which the data is addressed to all users itis called broadcasting.

One way of broadcasting data is to use an IP datacasting (IPDC) network.IPDC is a combination of digital broadcast and Internet Protocol.Through such an IP-based broadcasting network, one or more serviceproviders can supply different types of IP services including on-linenewspapers, radio, and television. These IP services are organized intoone or more media streams in the form of audio, video and/or other typesof data. To determine when and where these streams occur, users refer toan electronic service guide (ESG). One type of DVB is Digital VideoBroadcasting-handheld (DVB-H). The DVB-H is designed to deliver 10 Mbpsof data to a battery-powered terminal device.

DVB transport streams deliver compressed audio and video and data to auser via third party delivery networks. Moving Picture Expert Group(MPEG) is a technology by which encoded video, audio, and data within asingle program is multiplexed, with other programs, into a transportstream (TS). The TS is a packetized data stream, with fixed lengthpackets, including a header. The individual elements of a program, audioand video, are each carried within packets having an unique packetidentification (PID). To enable a receiver device to locate thedifferent elements of a particular program within the TS, ProgramSpecific Information (PSI), which is embedded into the TS, is supplied.In addition, additional Service Information (SI), a set of tablesadhering to the MPEG private section syntax, is incorporated into theTS. This enables a receiver device to correctly process the datacontained within the TS.

As stated above, the ESG fragments may be transported by IPDC over anetwork, such as for example, DVB-H to destination devices. The DVB-Hmay include, for example, separate audio, video and data streams. Thedestination device must then again determine the ordering of the ESGfragments and assemble them into useful information.

In a typical communication system, a cell may define a geographical areathat may be covered by a transmitter. The cell may be of any size andmay have neighboring cells. FIG. 3 illustrates schematically an exampleof cells, each of which may be covered by a different transmitter. Inthis example, Cell 1 represents a geographical area that is covered by atransmitter for a communication network. Cell 2 is next to Cell 1 andrepresents a second geographical area that may be covered by a differenttransmitter. Cell 2 may, for example, be a different cell within thesame network as Cell 1. Alternatively, Cell 2 may be in a networkdifferent from that of Cell 1. Cells 1, 3, 4, and 5 are neighboringcells of Cell 2, in this example.

In accordance with one or more embodiments, data used in channelsearches and service discovery is signaled using symbols at least in thebeginning of a data frame carrying multimedia and other data forservices. In other embodiments, one or more of these symbols may also beinserted within the data frames. Further, one or more of these symbolsmay be inserted at the beginning of, and/or within, a superframecomposed of two or more data frames.

In one embodiment, the symbols comprise a first symbol that can be usedfor identifying that the signal is of the desired type. Further, thefirst symbol may be used for detecting an offset from the radio channelcenter frequency. The symbols may comprise a second symbol that maycarry data relating to the modulation parameters that are used insubsequent data symbols. In another embodiment, the symbols comprise athird symbol that may be used for channel estimation.

FIG. 4 shows a frame and superframe of symbols, synchronization symbols,S1-S3, used for channel searches and service discovery, and data D inaccordance with an aspect of the invention.

In various digital broadcast networks, a multicarrier signal may bepositioned relative to the channel raster so that the signal centerfrequency (SCF) coincides with the channel center frequency (CCF), or itmay be offset from the channel center frequency. The signal centerfrequency may be offset due to frequency spectrum use reasons (e.g.interference from a neighboring channel). For the first symbol, not allavailable subcarriers are used. In various embodiments, the subcarriersthat are selected for the first symbol may be evenly spaced and may besymmetrically positioned with regard to the channel center frequency orthe offset signal frequency.

FIG. 5 shows how a signal center frequency may coincide with, or beoffset relative to, a channel center frequency (CCF). In FIG. 5, SCF Aand its corresponding CSF coincide, SCF B and SCF C are offset withregard to the corresponding CSFs. The rectangles in FIG. 5 illustratethe subcarriers selected for the first symbol from the availablesubcarriers. For SCF A, SCF B, and SCF C, the selected subcarriers arecentered around the respective SCFs. The selected subcarriers for SCF D,however, are centered around the CCF, as opposed to the SCF.

For the first symbol used for channel searches and service discovery,the subcarriers may be selected so that they may be found irrespectiveof the offset. In the first symbol, a fixed Fast Fourier Transform (FFT)may be used. The fixed FFT may be selected from the available FFT sizesthat in present digital video broadcast systems include 2K, 4K, 8K, butmay also include 1K at the lower end and 16K at the higher end. In oneembodiment, the lowest available FFT is used. Further, the first symbolmay use a fixed guard interval (GI) that may be selected from the GIsused for the symbols carrying data. The first symbol may, in oneembodiment, have no guard interval.

The number of subcarriers for the first symbol may be less than half ofthe available subcarriers.

When the first symbol is used for channel offset signaling, the carriersmay be modulated using Binary Phase Shift Keying (BPSK) or QuadraturePhase Shift Keying (QPSK). The selected pilot pattern may be differentfor different channel offset values, and the pilot pattern andsubcarrier modulation may be selected, in one embodiment, so that thedifferent pilot patterns are orthogonal to each other and differ fromeach other maximally thus allowing robustness in detection. In oneembodiment the different pilot patterns may signal the FFT-size only andthe frequency offset is found by detecting the shift from the nominalcenter frequency.

For the second (and third, if present) symbol the full signal bandwidth(substantially all available carriers) may be used. In an embodiment,the second (and third) symbol may use the same FFT size and guardinterval as the first symbol. In some embodiments, not all of theavailable subcarriers are used for the second (and third) symbols. Inone embodiment, the second and third symbol may have the samesubcarriers as pilot subcarriers and, in a further embodiment, haveadditional subcarriers used as pilots. In one embodiment, the secondsymbol also carries signaling data and, further, may carry forward errorcorrection data (FEC) for the signaling data.

In accordance with embodiments, a part of a signal (e.g.initialization/synchronization symbol(s)) is introduced that has knowncharacteristics and remains the same with different digital videobroadcast operation modes. The known part of signal contains parametersfor the rest of the signal; therefore, the rest of the signal can bedecoded without trial and error methods after the known part is decoded.Also, the channels containing digital video broadcast services can beefficiently detected using the known part of the signal. If the fixedknown part is not found from the examined signal, then the signal willbe considered a non-digital-video-broadcast signal or an empty channel,and the receiver can promptly proceed to a next channel/frequency.

FIG. 6 is a flow chart showing steps performed by a receiver inaccordance with at least one embodiment. A frequency synthesizer in thereceiver is programmed to the nominal center frequency of the channel,according to the channel raster, as shown at 602 for receiving a signalon the channel. An attempt is made to determine whether the receivedsignal is of a desired type and whether an offset is in use by comparingthe received signal to a stored set of known signals, as shown at 604.If a match is found, the signal is determined to be of the desired typeand the offset and FFT size for the signal can be determined. Adetermination is made with respect to whether a match was detected, asshown at 606. If a match is not detected, then the “no” branch from 606is followed, the channel is considered to contain anon-digital-video-broadcast signal or the received signal is not of thedesired type, and processing proceeds to the next channel, as shown at608.

Otherwise, if a match is detected, then the “yes” branch from 606 isfollowed, the determined frequency offset is used for reprogramming thefrequency synthesizer, as shown at 610. The next synchronization symbolis demodulated to detect modulation parameters for data symbols, asshown at 612. Finally, channel estimation and correction and datademodulation are then performed, as shown at 614.

In case the reprogramming of the frequency synthesizer takes arelatively long time, the receiver may wait for the next set ofinitialization/synchronization symbols and demodulate the modulationparameters from that set.

FIG. 7 shows an example of the size of a pilot signal bandwidth relativeto a signal bandwidth and a channel raster bandwidth in accordance withan aspect of the invention. In an embodiment, the first symbol is apilot symbol for coarse frequency and timing synchronization. Thebandwidth of the pilot symbol is smaller than the actual data symbol,e.g. in 8 MHz data symbol case, the pilot symbol could be 7 MHz wide.The pilot symbol center frequency may be the same as the frequency forthe data symbols, i.e., in case an offset is used for data symbols, theoffset may also be used for the pilot symbol. With the pilot symbol'ssmaller bandwidth, the receiver's RF part can be programmed to thenominal channel center frequency during the initial synchronizationphase and still be set to receive the pilot symbol's whole bandwidth.Without the pilot symbol's smaller bandwidth, the receiver's RF channelselection filter would filter out part of the pilot symbol.

In an embodiment, the pilot symbol may use known (fixed) FFT and GuardInterval selection. Also the number of used pilots may be different thanfor data symbols, i.e., part of the pilots can be extinguished, e.g.,256 pilots could be used. The pilots may be modulated with a knownsequence.

FIG. 8 illustrates sparse pilot spacing of a pilot sequence for a pilotsymbol in accordance with an aspect of the invention. The modulationsequence “finger print” for the pilot pattern may be known by thereceiver. In addition to modulation, the subcarriers in the pilotsymbols may also have different boosting levels, as illustrated in FIG.8.

FIG. 9 is a flowchart showing steps performed by a receiver forperforming correlation in the frequency domain to detect the coarseoffset being used. A radio frequency part of the receiver (frequencysynthesizer) is programmed to the nominal center frequency (according tothe channel raster) of the channel, as shown at 902.

An FFT is calculated using a predetermined FFT size as shown at 904. Thewidth of the pilot symbol is smaller than the channel bandwidth.Therefore, the FFT is able to capture the pilot symbol even when aninitial setting for frequency synthesizer is wrong because of theoffset.

The frequency offset is detected based on the offset of the pilotsynchronization symbol in the frequency domain, as shown at 906. Ifnon-correlation in the frequency domain is found, then the signal is nota digital video broadcast signal and channel search can proceed to thenext channel.

The offset is compensated for by reprogramming the receiver's frequencysynthesizer, as shown at 908. The next synchronization symbol isdemodulated to detect modulation parameters for data symbols, as shownat 910. Channel estimation and correction based on the channelestimation symbol is performed, as shown at 912, and then data isdemodulated as shown at 914. In an embodiment the receiver may wait fora synchronization symbol in the next set of synchronization symbols thusallowing the frequency synthesizer to be reprogrammed to the signalcenter frequency.

Different pilot sequences (finger prints) may be used based on theoffset in use. For example, if 7 offsets are possible (± 3/6 MHz, ± 2/6MHz, ±⅙ MHz, 0), 7 different pilot sequences may be introduced. Severalmethods can be utilized to construct the pilot sequence including, butnot limited to: pseudo random sequence, inverting every second, boostingthe center carrier, and the like. In accordance with an embodiment, thereceiver performs a correlation in the time domain to detect the usedpilot sequence, and, therefore, the offset used. The finger prints maybe used in accordance with one or more embodiments directed toperforming a time domain correlation. But, frequency domain embodiments,the offset can be detected by a sliding correlator in the frequencydomain, that is, a single finger print may be used. Additionally, onecould code information like FFT size for frequency domain embodiments ifdifferent finger prints are used for different FFT sizes, for example.Then, a frequency domain correlation could be run with several fingerprints. In an embodiment, if there are several finger prints in use, thereceived fingerprint may be compared simultaneously to several storedfinger prints. A received pilot sequence may be translated in frequencydomain stepwise over the channel bandwidth, wherein a high correlationsignal is produced when the pilot sequences coincide.

FIG. 10 is a flow chart that shows steps in accordance with anembodiment for performing a service discovery correlation in the timedomain. A radio frequency part of the receiver (frequency synthesizer)is programmed to the nominal center frequency (according to the channelraster) of the channel, as shown at 1002.

In one embodiment a correlation of the received pilot sequence isperformed in the time domain with known pilot sequences to detect theoffset used, as shown at 1004. For examples, if seven offsets are inuse, seven different pilot sequences (finger prints) are defined. Eachcoarse offset corresponds to a particular pilot sequence finger print.Based on the correlation, the finger print used, i.e., the offset used,can be detected. The pilot sequence will be in the nominal centerfrequency of the channel (according to the channel raster). In oneembodiment a set of pilot symbols are defined so that each of themcorresponds to a frequency offset-FFT size pair, wherein based on thedetected correlation both the offset and FFT size can be detected.

The frequency offset is detected based on the identified pilot sequencefinger print, as shown at 1006. If none of the pilot sequences showcorrelation, then the signal is not a desired digital video broadcastsignal, and search can proceed to the next channel.

The offset is compensated for by reprogramming the receiver's frequencysynthesizer, as shown at 1008. The next synchronization symbol isdemodulated to detect modulation parameters for data symbols, as shownat 1010. Channel estimation and correction based on the channelestimation symbol is performed, as shown at 1012, and then data isdemodulated as shown at 1014. In one embodiment the receiver may waitfor a next set of synchronization symbols for allowing the frequencysynthesizer to be reprogrammed.

After the offset has been found and the frequency synthesizer isreprogrammed, the second symbol (i.e., the symbol following the pilotsymbol) may use fixed FFT and Guard Interval selection, but would usethe full signal bandwidth. The second symbol may then contain specificinformation about the modulation parameters for the subsequent datasymbols. In another embodiment the second symbol may use the FFT that issignaled in the first symbol.

An optional third symbol could be inserted before the data symbols tofacilitate channel estimation.

FIG. 11 shows an example of a pilot/signaling symbol sequence inaccordance with an aspect of the invention. The pilot symbol 1102 andthe Signaling Symbols 1104 and 1106 may be repeated in the transmissionfrequently enough, e.g., every 50 ms, to enable signal detection andsynchronization as fast as is desired. The first pilot symbol 1102 isused for coarse frequency and time synchronization, and, in addition, itmay also carry information on the FFT size for the following symbols.The FFT, guard interval, and the modulation are fixed for the firstsymbol. In one embodiment the second symbol 1104 comprises the samepilot subcarrier as the first symbol but may have, in addition, moresubcarriers that are used as pilot subcarriers. The second signalingsymbol also carries signaling data comprising FFT size, guard interval,and modulation parameters. The third signaling symbol comprises stillmore pilots that are used for channel estimation and fine timing.

The modulation parameter for data symbols (like constellation, QPSK vs.16QAM vs. 64QAM) may be varied frequently because the repeated signalingsymbols carry information about the selected parameters.

FIG. 12 is a flowchart showing steps of a method performed by atransmitter in accordance with at least one aspect of the invention. Asymbol sequence is composed that includes a pilot symbol configured toconvey coarse frequency and timing synchronization information as thefirst symbol followed by a next signaling symbol configured to conveymodulation parameters as the second symbol, which is followed by aplurality of data symbols, as shown at 1202. In one embodiment thesecond signaling symbol may be followed by a third signaling symbol. Thesymbol sequence is then transmitted on a broadcast channel with apilot-signal bandwidth that may be narrower than a data-signalbandwidth, which further may be narrower than a channel raster bandwidthof the broadcast channel, as shown at 2004.

FIG. 13 and FIG. 14 depict the relation between P1, P2 and DATA symbols(i.e. OFDM symbols) by example. From FIGS. 13 and 14, it may be seen howdata has been split for the duration of P2 and data symbols. The datapackets may be placed immediately after the last P2-n packet and bothare carried within the ‘DATA symbols.’

FIG. 15 shows an exemplary frame and slot structure in accordance withat least one aspect of the invention. In FIG. 15, a frame 1502 mayconsist of one or more slots 1504. For example, frame 1502 includes slot1 1506 through slot 4 1512. Each slot 1506-1512 may include several OFDM(orthogonal frequency division multiplexing) symbols, typically from afew symbols up to some tens of symbols. The services are allocated tothese slots so that one or more slots are used for a service. Forexample, slot 1 1506 may includes a number of OFDM symbols 1514 through1524. Furthermore, each OFDM symbol may include numerous OFDM cells. Forinstance, OFDM symbol 1514 includes OFDM cells 1526 through 1534.

Embodiments are related to initial service discovery in a Digital VideoBroadcasting-Terrestrial next generation (DVB-T2) system. The DVB-T2system may include a preamble, which is intended for efficientidentification of available T2 signals. The preamble should not consumetoo much capacity, but it should be compatible with different FastFourier Transform (FFT) sizes (2 k, 4 k, 8 k, 16 k, and 32 k).Minimizing the overhead has lead to using a 2 k symbol (P1) for eachFFT-size and signalling the actual FFT-size within this symbol bymodulating the carriers by different pseudo random binary sequences(PRBS). To find out the FFT-size of the following symbols, the receiverdetects the modulating PRBS. This PRBS also indicates the integerfrequency shift (DVB-T2 signal can be shifted by +/−⅙, +/− 2/6, +/− 3/6MHz compared to the nominal center frequency). To summarise, the P1symbols are used in the initial scan to: (1) detect the presence of T2signal; (2) estimate the frequency offset; and (3) detect the used FFTsize.

After the initial scan, the P1 symbol may not be used during normal datareception or handover because the parameters carried by P1 (i.e., FFTsize and frequency offset) remain constant. With respect to handovers,these parameters are the same between radio frequency (RF) channels orthey are signalled before a handover (e.g., in Program SpecificInformation/Service Information (PSI/SI) according to ETSI EN 300 468Digital Video Broadcasting (DVB); Specification for Service Information(SI) in DVB systems). P1 can, however, be used during normal datareception to, for example, detect the frame start or to improvesynchronization and channel estimation algorithms. P2 symbol(s) is/are asignalling and channel estimation symbol(s) which is/are located afterthe P1.

The detection of P1, and, thus, the detection of the DVB-T2 signal, isbased on a guard interval correlation (GIC). In GIC, the guard intervalis correlated with the end of the symbol. A peak in the GIC indicates apotential DVB-T2 signal, which may be verified from the P2 symbol. Thefirst problem is that the guard interval should be long in order toprovide robust detection (i.e., a long guard interval provides highersignal to noise ratio). A longer guard interval, and thus a longer P1,however, decreases data capacity.

Since P1 is the first symbol to be received, there is typically no priorknowledge of the channel conditions. Therefore, the P1 symbol shouldinclude some means to overcome the channel distortions. In practice,this would mean using, e.g., extra pilot carriers for channel estimationor differential modulation between the subcarriers.

Because of the lower FFT-size, the carrier spacing of the P1 symbol maynot be as dense as in the following data symbols (e.g., 2 k for P1 and32 k for data). For a successful PRBS detection in P1, the coherencebandwidth of the channel should be smaller than the subcarrier spacingof a 2 k symbol. However, the network might be designed for the 32 kmode, and the long Single Frequency Network (SFN) delays might producemuch higher frequency selectivity.

The complex valued received signal at carrier index k may be expressedas r_(k)=h_(k)s_(k)+n_(k), where s_(k) is the transmitted data symbol(e.g. using Quadrature Phase Shift Keying (QPSK)), h_(k) is the channelresponse at the carrier index k, and n_(k) is the noise term.

In coherent demodulation, h_(k) is first estimated using pilots, andthen the effect of the channel is equalized by, for example, dividingr_(k) by estimated h_(k).

If we consider DVB-T2 and P1 symbol, there are no pilots to estimateh_(k). Therefore, non-coherent demodulation will typically be used,without channel estimation. This may be done by using differentialmodulation (e.g. Differential Binary Phase Shift Keying (DBPSK)) wherethe information is coded to the phase difference between two adjacentcarriers. These two adjacent carriers may be expressed asr_(k)=h_(k)s_(k)+n_(k) and r_(k+1)=h_(k+1)s_(k+1)+n_(k+1). Thetransmitted symbol may be decoded from the phase difference betweenthese two received carriers: r_(k+1)−r_(k)=h_(k+1)s_(k+1)−h_(k)s_(k)+n.

FIG. 16 illustrates coherence bandwidth and differential modulationwithin one pilot (P1) OFDM symbol. It is assumed that the phases of thechannel responses h_(k) and h_(k+1) are approximately the same, as shownin the upper graph of FIG. 16. However, in a highly frequency selectivechannel (e.g., the lower graph of FIG. 16), the correlation betweenadjacent channel responses is relatively low. This makes itimpracticable to use differential modulation between carriers.

The coherence bandwidth (i.e., the bandwidth where the channel responseis highly correlated) can be approximated by

${W_{coh} \approx \frac{1}{\tau_{d}}},$where τ_(d) is the delay spread of the channel. The coherence bandwidthof the channel should be lower than the carrier spacing in order to useDBPSK between carriers. The FFT size of P1 is 2 k, and the carrierspacing in 8 MHz channel is 4.46 kHz. From these carriers every 3^(rd)or 9^(th) carrier is used. Therefore, the actual carrier spacing can beeven 40.1 kHz. On the other hand, the delay spread in large SFN networkcan be 448 μs (16 k mode with ¼ guard interval) resulting in a coherencebandwidth of 2.2 kHz.

In accordance with an aspect of the invention, two P1 symbols are used,e.g., 1 k symbol with GI=1/1. Both symbols are separately used in GIC.When GI=1/1, the whole symbol duration may be utilized in GIC.

In accordance with an aspect of the invention, differential modulationis applied between two P1 symbols as shown in FIG. 17. Because thedifferential modulation is now performed subcarrierwise, there are norequirements for the coherence bandwidth. (Alternatively, the first P1symbol could be used for channel estimation, which would allow coherentdemodulation for the second P1 symbol.)

The time interval of two P1 symbols is relatively short such that thechannel does not change from the first symbol to the second symbol.Therefore, in accordance with one or more embodiments, the differentialmodulation can be done in the time domain between carriers having thesame carrier number.

Embodiments also support mobile reception. In accordance withembodiments, the coherence time of the channel is longer than theduration of two P1 symbols. This makes the correlation between r_(k)(1)and r_(k)(2) high. The coherence time of the channel can be approximatedby

${\tau_{coh} \approx \frac{1}{F_{d}}},$where F_(d) is the Doppler spread of the channel and it is given by

${F_{d} = {\frac{v}{c}F_{c}}},$where v is the speed of the receiver, c is the speed of the light(3*10^8 m/s), and F_(c) is the carrier frequency. If v=120 km/h andF_(c)=666 MHz, then F_(d)=74 Hz and τ_(coh)=13.5 ms, which issignificantly longer than the duration of a P1 symbol (e.g., 280 μs).

In accordance with one or more embodiments, symbol synchronization of P1may be improved. The P1 symbols may have a 1/1 guard interval, whichwould improve symbol synchronization and maximize guard intervalcorrelation length with respect to overhead. The P1 symbols may use a 1k FFT, which would decrease overhead compared to two 2 k symbols.

Guard interval correlation (GIC) is a basic method for synchronizationin Orthogonal Frequency Division Multiplexing (OFDM) symbols. Since theGI is a cyclic copy of the last part of the actual OFDM symbol, thereceiver is able to find the beginning of an OFDM symbol by detectingthis correlation. In practice, the receiver continuously correlates twoblocks of the received signal, which are separated by N samples (N isthe FFT size and also the number data samples). A correlation peak isdetected at the correct position.

FIG. 18 shows two 1 k symbols with 1/1 guard interval and differentialmodulation between the symbols. As can be seen, the 1/1 guard intervalmeans that the GI and the data part have the same length, and thesamples are also equal. Equivalently, the 1/1 symbol could be said tohave two equal symbols without guard interval.

Because of differential modulation, the consecutive symbols, P1 and P1′,are different, which means that a normal GIC should be applied withineach P1 symbol. The correlation length, however, is doubled compared toa 2 k 1/4 GI symbol (¼*2048=512), and the correlations from the twosymbols can be combined for further improvements. The 1 k 1/1 GI symbolis also desirable since the guard interval correlation does not nowmatch to the data modes (2 k, 4 k, etc.).

Another embodiment speeds up the initial scan. It is desirable toquickly detect non-T2 signals so that the receiver can tune to the nextfrequency. This can be done by detecting the zero carriers in the P1symbol by: (1) calculating three sums (see FIG. 19, which showscalculation of sums of received energy from P1 in accordance with one ormore embodiments) of received energy for carriers that belong to thesubsets r_(3k), r_(3k−1), and r_(3k+1), where r is the kth carrier of P1symbol(s) and k=1, 2, 3 . . . ; and (2) detecting the existence ofT2-signal by comparing the received energy on the three subsets; and (3)setting an energy threshold (e.g. 5 dB below the strongest); and (4) ifonly one sum exceeds the threshold, a possible T2 signal is detected.

FIG. 20 shows a transmitter in accordance with one or more embodiments.The first P1 is BPSK modulated according to a reference sequence, andthe second P1 is modulated as follows: if PRBS_(k)=0→b_(k,2)=b_(k,1); ifPRBS_(k)=1→b_(k,2)=−b_(k,1) (or vice versa), where PRBS_(k) is the kthelement of the PRBS, and b_(k,m) is the transmitted symbol on kthcarrier at mth P1 symbol. Next, the transmitter combines the originalreference sequence and the delayed differentially modulated sequencebefore the Inverse Fast Fourier Transform (IFFT) and guard intervalinsertion. N refers to the FFT size.

FIG. 21 shows a receiver in accordance with one or more embodiments. Thereceiver performs the inverse of the transmitter operations discussedabove in connection with FIG. 20. That is, the receiver removes theguard interval from the P1 symbols (first and second pilot symbol),performs a fast fourier transform on the P1 symbols, and thereafterdifferentially demodulates P1 symbols to obtain an estimate of thetransmitted pseudo random binary sequence. The receiver does not have toknow the reference sequence.

FIG. 22 is a flow diagram that shows steps that may be performed by areceiver in accordance with one or more embodiments. In the initialscan, the receiver may be tuned to the nominal centre frequency of thechannel, and it may start to look for the P1 symbol. The followingprocedure may then be repeated at selected channels (and bandwidths)—butnot necessarily with every frequency offset since the P1 symbol may bedetected at the nominal centre frequency regardless of the used offset.

The first task after bandwidth and nominal centre frequency selection isto find the existence of a T2-signal. The P1 symbol can be found, e.g.,by guard interval correlation, which is immune to frequency offset.Using guard interval correlation helps also in T2-signal detection sincethe lack of a 2 k symbol implicates a non-T2 channel.

Guard interval correlation is intended for situations where the delayspread of the channel stays within the guard interval, which may not bethe case with P1 symbol in large scale SFNs (e.g. with 32 k mode). Inthis case, delays longer than the guard interval—especially delays thatare multiples of the useful symbol duration—produce false correlation.

It should be noted, however, that the symbol timing in presence ofstrong SFN echoes is not only a P1-specific problem because the receiveranyway should be able to synchronize to the correct path. The differenceis that P1 correlation has higher noise level because of the shorter GICwindow.

Coarse time and fractional frequency synchronization are obtained fromthe guard interval correlation. These are coarse estimates that are usedfor the P1 symbol itself, and they may be refined using the followingsymbols. It is assumed that these estimates are accurate enough todetect one of the five PRBS patterns to find the FFT-size.

For a fast initial scan, the channels that do not contain a T2-signalshould be discarded relatively quickly. The preamble structure inaccordance with embodiments supports a stepwise detection where thenon-T2 channels can be discarded relatively quickly, and the detectionof a T2-signal may be confirmed by reading the L1 static signalling.

First elimination may be done by the guard interval correlation. P1signal may be repeated every frame (around 200 ms) and it is ratherrobust in terms of SNR requirements so testing two consecutive P1positions may be reliable enough to detect the T2 signal. This wouldtake around 500 ms per RF-channel. A receiver may then decide whether apossible P1 symbol has been found. If this is done over the 39UHF-channels and even with 3 channel bandwidths, the total time used forthe scan is approximately 58 seconds. Note that trying to scan differentbandwidths at the same time does not really help as the channel rastersare different.

Once a possible P1 symbol has been found, the receiver may performcoarse synchronization and FFT. Next, the receiver may use the sparsecarrier raster to differentiate between T2- and other 2 k signals. Thus,the non-T2 signals can most likely be detected from the first receivedP1 symbol.

Detection of the frequency offset is based on finding the shifted pilotpattern. The detection of frequency offset and FFT-size may be separatedby first using the power at the assumed pilot carriers to find thecorrect offset, and, after that, calculating the correlation to the fivePRBSs. On the other hand, the PRBSs could be used already when findingthe frequency offset. The sparse carrier raster decreases the complexityof the search algorithms.

After the frequency offset has been detected, the receiver can be tunedto receive the data symbols. Another task is to find out the used guardinterval to decode the P2 symbol. As the P1 symbol does not carry anysignalling information of the GI, the receiver may detect this by usingthe normal OFDM symbols during the frame. The P2 symbol immediatelyafter the detected P1 can not be decoded. But there is enough time todetect the GI before the next frame as the whole 200 ms frame durationcan be used. This adds another 200 ms to the signal acquisition time,but this happens most likely only with the FOUND T2-signals, not withevery tested channel. As the maximum number of parallel multiplexestypically is in the order of 7 to 8, the total time added to the scansequence is less than 2 s.

In case the frame duration is configurable, the frame synchronizationcan be obtained by recognising the next P1 symbol. The detectedparameters from the L1 static signalling in the P2 symbols are thenconfirmed.

In one embodiment, the first P1 is used for a channel estimate, which isthen used to equalize the second P1. This re-uses an underlying idea ofvarious embodiments although the implementation is different. N refersto the FFT size.

In accordance with DVB-T2 standards, P1 and P2 symbols are presented asa solution for initial scan and transmission of signalling. Inaccordance with embodiments, differential modulation between two P1symbols may have advantages in highly frequency selective channels.

As discussed above, the P1 symbols are used in the initial scan to: (1)detect the presence of T2 signal; (2) estimate the frequency offset; and(3) detect the used FFT size. A possible method estimating the frequencyoffset (and to some extent detecting the presence of T2 signal) is touse a frequency domain ‘comb’, i.e., use a subset of the availablesubcarriers in the OFDM symbol. Assume that there are a total of Lsubcarriers available (=FFT-size with guard bands deducted). Further,assume that every third subcarrier is available for thispilot/synchronization usage, so there will be L′=└L/3┘+1 activesubcarriers for the synchronization signal. Mathematically, the comb maybe represented with a sequence P(0), (P1), . . . , P(L′−1) of bits.Here, the bit P(k) tells, whether subcarrier number lowest+3*k containsa Binary Phase Shift Keying (BPSK) signal or not: ‘0’ indicates asubcarrier containing no power, and ‘1’ indicates a subcarriercontaining a BPSK-modulated signal. The idea is that when the operatoruses a channel frequency offset, the comb is shifted accordingly. Thus,after achieving timing synchronization and fractional frequencysynchronization, the receiver may carry out FFT and search for theinteger frequency offset. Here the receiver may use the received powerat the presumed pilot carriers (i.e. the comb) and find the frequencyoffset without demodulating the pseudo random binary sequence. Thecorrect integer frequency offset (=an integer multiple of subcarrierspacing) can then be detected by the presence of a relatively good matchwith the shifted comb and the measured subcarrier signal power. TheFFT-size (selected from, for example, 5 choices) is then indicated by aselection of 5 BPSK-patterns S_(m)(0), S_(m)(1), . . . S_(m)(L′−1), form=1, 2, 3, 4 or 5.

The frequency offset (after adjusting for its fractional part) amountsto adding a constant offset n to the subscripts. The sum

${S(n)} = {\sum\limits_{k = 0}^{L - 1 - n}{{P(k)}{P\left( {k + n} \right)}}}$then computes the number of collisions between the comb and its shiftedversion, and S(0)=N equals the number N of subcarriers in the comb. Fordetection of the integer frequency offset to work, the collision countsS(n), n≠0 should be small in comparison to the correct match N.

Ideally, the structure of the P1-signals should be such that it supportsother methods of detection as well, thereby providing hardware designerswith freedom of choice. Another approach to the problem of detecting thepresence of a P1-signal is based on time domain correlation. In order tosupport this alternative approach as well, the actual signals

$\sum\limits_{k = 0}^{L - 1}{{P(k)}{S_{m}(k)}{\exp\left( {2\;\pi\;{j\left\lbrack {f + {\left( {n + {3\; k}} \right)\Delta\; f}} \right\rbrack}t} \right)}}$should have good cross-correlation properties—not only for distinctvalues of m but for distinct values of (m,n) pairs, i.e., for differentvalues of the (FFT-size, frequency offset) combinations.

Other properties required by the set of signals are reasonable timedomain autocorrelation properties and reasonable peak-to-average-powerratio (PAPR) properties. Ideally, it should also be possible to quicklyand efficiently regenerate both the comb and the BPSK-sequences withoutresorting to large look-up tables.

Embodiments are directed to: 1) combs limited to every third subcarrier,and 2) combs that contain approximately one half of the remainingsubcarriers, so the number of active subcarriers N should beapproximately L/6. With these assumptions in place, shorter combpatterns/sequences of length L′=└L/3┘+1 are of interest.

In accordance with embodiments, a binary m-sequence of a suitable lengthis used to generate the comb, and selected cyclic shifts of the samem-sequence (now interpreted as +1/−1 as opposed to 0/1) are used togenerate 5 BPSK-patterns.

Six bit patterns, each consisting of r bits, not all zero, which arehereafter referred to as the seeds, are specified. The seeds are thenextended into a sequence of length 2^(r)−1 by applying a recurrenceformula determined by a primitive polynomial of degree r. Note that thesame recurrence formula is applied to form each of the 6 sequences. Oneof the sequences is singled out to determine the comb, and the other 5determine the BPSK-patterns by reinterpreting ‘0’ as +1 and ‘1’ as −1.Ideally, then L′=2^(r)−1. Different use cases, and an alternative methodfor constructing the comb, may also be used.

In the specific use case of DVB-T2, L=1531 subcarriers, so L′=511=2⁹−1,r=9, and the primitive feedback polynomial 1+x⁴+x⁹ may be used. Anexample set of seeds consists of 100 000 000 for the comb, and 000 110101, 110 001100, 101 111 101, 101 101 111, 111 100 111 (all interpretedas +/−1 s) for the 5 BPSK-patterns. These are extended to sequences P,and S_(m) for m=1, 2, 3, 4 and 5 by repeated applications of therecurrence formulas and P(k)=P(k−4)+P(k−9) (mod 2),S_(m)(k)=S_(m)(k−4)*S_(m)(k−9), for k=9, 10, . . . , 510.

A design criterion in the selection of the seeds is that, while theresulting sequences are cyclic shifts of one another, the amount ofshift that it takes to get from one to another should be made relativelylarge. Similarly, the seeds may be designed such that one of them cannotbe produced from the bitwise XOR of the comb sequence and anothersequence by a short (e.g., less than 45 positions) cyclic shift.

If the number of available carriers L′ is not of the form 2^(r)−1, butis still relatively close to such a number, then the comb and thesequences may be shortened by truncating a small segment off the tailend of the m-sequences, or the pattern may be extended by cyclicallyrepeating it for a relatively short time. In the above example, thenumber of subcarriers may be reduced from 1531 to 1507 by cyclicallyshifting the comb pattern as well as the BPSK-sequences by one position.To achieve this, the 9 bit seeds may be extended to 10 bits by applyingthe recurrence relation once. After that, the first bit may be left outthereby producing a 9 bit seed. Thus, the seed 000 000 001 for the comb,and the seeds 001 101 010, 100 011 000, 011 111 010, 011 011 110, 111001 111 for the BPSK-sequences would be used instead of the abovesuggestions. Then, the comb will begin with 8 zeros, i.e. 24 emptysubcarriers, and the P1-signal is narrowed down to 1507 consecutivecarriers. Observe that the role played by the available bandwidth isless important, as in a narrower band (e.g. 5 MHz) application thespacing between subcarriers is also narrower, and there is still roomfor roughly the same number of subcarriers.

An alternative method of generating a frequency domain comb is to usethe quadratic residue sequences (=QR-sequences), which are known in theart. The resulting comb shares the collision statistics between shiftedversions with the m-sequence based comb. This alternative method has theadvantage that the length of a QR-sequence is a prime number p congruentto 3 modulo 4. Thus, the set of available lengths is more flexible whenQR-sequences are used. Cyclically shifted versions of the same sequencecan be used also here for constructing the BPSK-sequences. However,generating a relatively long QR-sequence on the fly is computationallymore taxing, and in practice a relatively large look up table may haveto be used.

In accordance with at least one embodiment, the proposed 5 P1-signalsare

${{P\; 1_{m}(t)} = {\sum\limits_{k = 0}^{510}{{P(k)}{S_{m}(k)}{\exp\left( {2\;\pi\;{j\left\lbrack {f + {\left( {n + {3\; k}} \right)\Delta\; f}} \right\rbrack}t} \right)}}}},$for m=1, 2, 3, 4 and 5. Here n denotes the integer part of the frequencyoffset. It is counted as a multiple of subcarrier spacing, so in theproposed use case n=±37, ±75, ±112 correspond to frequency offsets of±⅙, ±⅓, ±½ MHz (note that fractions of subcarrier spacing are handledearlier irrespective of whether they are a result of a rounding errorhere or a result of a clock discrepancy between the receiver and thetransmitter). But the presented construction actually allows any integervalues of n up to 134. Here P and S_(m) for m=1, 2, 3, 4 and 5, are thesequences of length 511 discussed above. These signals occupy 256subcarriers within a range of 1531 consecutive subcarriers.

There are various other choices for the seeds that work equally well.For example, each of the 6 m-sequences may be cyclically shifted by thesame amount without changing the correlation properties. The examplevalues of seeds work well, when the integer part n of the frequencyoffset is less than 3*45=135. Within that range the cross-correlationsamong the offset versions of the sequences remain low. A computer searchhas revealed other sets of seeds with equally good performance. Thepossibility of an even slightly wider range of low correlation has notbeen fully excluded, but it is known that if n might be as large as3*51=153, such a low correlation range cannot be achieved with thismethod, no matter how carefully the seeds are selected.

The spacing in multiples of 3 allows the integer part of the frequencyoffset to be detected relatively quickly as there are no collisionsbetween the true comb and the tested version, unless the differencebetween the tested and the actual integer offsets is a multiple of 3. Ifthat condition is satisfied, then the number of collisions is 256, whenwe have the correct offset, and the number of collisions is in the range119 . . . 128 otherwise, i.e. near the optimal pseudorandom half-waypoint of 128. For a randomly generated comb with a similar structure(=limited to every third subcarrier) and density (=altogether one out of6 subcarriers in the average is ‘active’), the expected range (+/−2standard deviations from the expected value) for the number ofcollisions is from 104 to 144, so the use of m-sequences improves thisby bringing the variation into a narrower range.

The underlying algebraic structure of the m-sequences helps ensure thatnearly all the sequences produced in this manner have reasonably goodPAPR-properties (the exception is to use the same seed for the comb andone of the sequences) and reasonably good time domain autocorrelations.The careful choice of the seeds further helps ensure goodcross-correlation properties among the offset version of the varioussequences. Indeed, the non-trivial correlations are very close to zeroas opposed to randomly fluctuating up to +/−2SD level of 32.

FIG. 23 is a graph of auto/cross-correlations between pilot sequencesand their frequency offset versions in accordance with one or moreembodiments.

FIG. 24 is a zoomed in version of the graph of FIG. 23 showing the lowcross-correlation range of frequency offsets.

FIG. 25 is a graph that shows the envelope amplitude of a firstP1-signal (a single symbol computed with a central frequency of 666 MHz,and carrier spacing of 4464 Hz, sampled at 25 MHz to produce thesefigures) in accordance with at least one embodiment. The scale is chosenso that the mean square amplitude equals one.

FIG. 26 is a zoomed in version of the graph of FIG. 25. Together FIGS.25 and 26 show the reasonable PAPR-properties of the set.

In the discussion of the BPSK and P1 sequences that follows, F=GF(512)will denote the finite field of 512 elements, and g will be a primitiveelement of F that satisfies the equation 1+g⁵+g⁹=0, so the powers g^(i)go through the non-zero elements of F, as the exponent i takes thevalues i=0, 1, . . . , 510. We further note that g⁻¹ will then be a rootof the earlier feedback equation 1+x⁴+x⁹=0. Let tr:F→GF(2) be the tracefunction. The previous 0/1-valued m-sequence and all its cyclic shiftsare gotten as the sequences m_(α)(i)=tr(αg^(i−1)), for i=1, 2, . . . ,511 and α≠0. We write e(x)=(−1)^(tr(x)), and ω=e^(2πj/511). Thus, we canselect elements αεF and β_(j)εF, j=1, 2, 3, 4, 5 such that the comb ofzeros and ones is gotten as P(i)=tr(αg^(i−1))=(1−e(αg^(i−1)))/2, andthat the BPSK-sequences are gotten as S_(j)(i)=e(β_(j)g^(i−1)). TheP1-sequences are thus given by the formulaP1_(j)(i)=(1e(αg^(i−1)))e(β_(j)g^(i−1))/2.

We have the identity e(x±y)=e(x)·e(y) and sums

${S(\gamma)} = {{\sum\limits_{x \in F}{{\mathbb{e}}\left( {\gamma\; x} \right)}} = 0}$(hereafter referred to as equation (1) or sum (1)), whenever γ isnon-zero, and the so called Gauss' sums

${S\left( {k,\gamma} \right)} = {\sum\limits_{i = 0}^{510}{{{\mathbb{e}}\left( {\gamma\; g^{i}} \right)}\omega^{kj}}}$(hereafter referred to as equation (2) or sum (2)) that have complexabsolute value √{square root over (512)} when both γ and k are non-zero,and less than that when one but not both of them is zero.

At this point we record that the proposed comb corresponds to the choiceα=1.

Let us consider the number of collisions between the patternP(k)=(1−e(αg^(k)))/2 and its shifted version P(k+n), where n indicatesthe shifted amount (at most └112/3┘=37). If we were to continue thepattern of this comb cyclically with a period of 511, then the number ofcollisions may be computed. Denote the variable x=g^(k), and adopt theusual convention that F* is the set of non-zero elements in the field F.Then the number of ‘collisions modulo 511’ is (so k+n is computed modulo511)

${\sum\limits_{k = 1}^{511}{{P(k)}{P\left( {k + n} \right)}}} = {\frac{1}{4}{\left( {{\sum\limits_{x \in F^{*}}1} - {\sum\limits_{x \in F^{*}}{{\mathbb{e}}\left( {\alpha\; x} \right)}} - {\sum\limits_{x \in F^{*}}{{\mathbb{e}}\left( {\alpha\; g^{n}x} \right)}} + {\sum\limits_{x \in F^{*}}{{\mathbb{e}}\left( {{\alpha\left( {1 + g^{n}} \right)}x} \right)}}} \right).}}$

Here the first sum is 511. Because t<511, the coefficients α, αg^(n),α(1+g^(n)) are non-zero, and equation (1) tells us that the remainingsums are all equal to −1 (adjusting for the fact that the term e(0)=1 ismissing from the sums). Altogether we get that the shifted comb has512/4=128 collision with the cyclically extended comb. When we take thetail effects due to the sum k+n overflowing >511 into account, we see anexpected drop on the number of collisions. At n=1, 2, 3, 4, 6, 7, 8there are 128 collisions, and this number drops approximately linearlyas n grows. When n reaches the maximum value of 37, the number ofcollisions is 125. The lowest value of 119 collisions is reached withoffset n=36. So with this comb the number of collisions between twooffset combs will be close to the ideal midway point of 128.

We can compute the cross-correlation between two P1-sequences (in thef-domain, as per Parseval's theorem it doesn't matter whether this isdone in the frequency or time domain) as

$\begin{matrix}{\left\langle {{P\; 1_{j}},{P\; 1_{j^{\prime}}}} \right\rangle = {\frac{1}{2}{\sum\limits_{i = 0}^{510}{\left( {1 - {{\mathbb{e}}\left( {\alpha\; g^{i}} \right)}} \right){{\mathbb{e}}\left( {\left( {\beta_{j} - \beta_{j^{\prime}}} \right)g^{i}} \right)}}}}} \\{{= {\frac{1}{2}\left( {{\sum\limits_{x \in F}{{\mathbb{e}}\left( {\left( {\beta_{j} - \beta_{j^{\prime}}} \right)x} \right)}} - {\sum\limits_{x \in F}{{\mathbb{e}}\left( {\left( {\alpha + \beta_{j} - \beta_{j^{\prime}}} \right)x} \right)}}} \right)}},}\end{matrix}$so the sum (1) tells us that this cross-correlation is equal to zero,provided that β_(j)−β_(j′) is non-zero (in other words, the twosequences are different) and that α+β_(j)−β_(j′) is non-zero (in otherwords, the two sequences are not bitwise complements of each other). Apractical test for this is that for two sequences of this type to beorthogonal, their initial segments are different from each other, andthat the bitwise XOR of their initial segments differ from the initialsegment of the comb P.

As in the computation of the number of collisions, we first extended thesequence cyclically in the f-domain, compute the cross-correlationbetween such an extended pair of signals, and more or less ignore theshort ‘tail’, which is the sum of a handful pseudorandom terms, and willnot contribute much. So the (f-domain) cross-correlation between aP1-signal and another P1-signal offset by t positions from the former is

$\begin{matrix}{\left\langle {{P\; 1_{j}},{P\; 1_{j^{\prime}}\left( {{offset}\mspace{14mu}{by}\mspace{14mu} n} \right)}} \right\rangle = {\frac{1}{4}{\sum\limits_{i = 0}^{510}{\left( {1 - {{\mathbb{e}}\left( {\alpha\; g^{i}} \right)}} \right){{\mathbb{e}}\left( {\beta_{j}g^{i}} \right)}}}}} \\{{\left( {1 - {{\mathbb{e}}\left( {\alpha\; g^{i + n}} \right)}} \right){{\mathbb{e}}\left( \beta_{j^{\prime}g}^{i + n} \right)}} =} \\{= {\frac{1}{4}\begin{pmatrix}{{\sum\limits_{x \in F}{{\mathbb{e}}\left( {\left\lbrack {\beta_{j} - {g^{n}\beta_{j^{\prime}}}} \right\rbrack x} \right)}} -} \\{{\sum\limits_{x \in F}{{\mathbb{e}}\left( {\left\lbrack {\left( {\alpha + \beta_{j}} \right) - {g^{n}\beta_{j^{\prime}}}} \right\rbrack x} \right)}} -} \\{{\sum\limits_{x \in F}{{\mathbb{e}}\left( {\left\lbrack {\beta_{j} - {g^{n}\left( {\alpha + \beta_{j^{\prime}}} \right)}} \right\rbrack x} \right)}} +} \\{\sum\limits_{x \in F}{{{\mathbb{e}}\left( {\left\lbrack {\left( {\alpha + \beta_{j}} \right) - {g^{n}\left( {\alpha + \beta_{j^{\prime}}} \right)}} \right\rbrack x} \right)}.}}\end{pmatrix}}}\end{matrix}$(hereafter referred to as equation (3)).

Observe that here the indices j and j′ may be equal, i.e., we are alsointerested in the correlation between a sequence and its offset version.From equation (1), we see that this main term is zero, unless one of thecoefficients in square brackets is zero. As n takes values in a rangeabout zero, we are left with the goal of selecting the coefficients β₁,. . . , β₅ in such a way that the base g discrete logarithms of thecoefficients themselves, and also of the sums α+β₁, . . . , α+β₅ are asfar from each other as possible (cyclically modulo 511). As there are 10field elements here altogether the minimum separation among the discretelogarithms cannot be higher than ℑ511/10┘=51. With the choice α=1=g⁰ ofthe sample construction a small heuristic search gave the set used inthe discussion above: β₁=g³³, α+β₁=g¹⁸¹, β₂=g¹³⁵, α+β₂=g⁴⁹⁹, β₃=g²⁴⁵,α+β₃=g³⁹⁸, β₄=g³⁴⁹, α+β₄=g⁸⁵, β₅=g⁴⁴⁵, α+β₅=g²⁹⁶. Here the discretelogarithms form a list {33, 135, 245, 349, 445, 181, 499, 398, 85,296}—the first five discrete logarithms specify the elements β₁, . . . ,β₅, and the last five list the discrete logarithms of the elements α+β₁,. . . , α+β₅. The smallest cyclic separation of 45 is here between 499and 33, as 33−499+511=45. Another sequence of discrete logarithms thatalso has the smallest cyclic separation of 45 is {33, 135, 233, 339,447, 181, 499, 388, 286, 80}. It is unknown, whether there are choicesleading to an even larger cyclic separation. As 3*45=135 (subcarrierseparations) is larger than 112, this suffices for our purposes.

These numbers explain the gaps in FIG. 23. There are no matches withoffsets up to 44 in either direction, so the width of the near-zero zonein FIG. 28 is 2*44+1=89 carriers. We note that the separation 45corresponds to the terms with a minus sign in equation (3). The smallestcyclic separation corresponding to a term with a plus sign is 96, and itoccurs between the pairs (445, 349) and (181, 85). This explains why thenearest sidelobes are all negative, and also explains the wider gap of2*96+1=193 carriers above the x-axis.

Here we show how the bound

${S\left( {k,\gamma} \right)} = {\sum\limits_{i = 0}^{510}{{{\mathbb{e}}\left( {\gamma\; g^{i}} \right)}\omega^{kj}}}$means that the auto-correlation of the proposed signals remains at a lowlevel at least for a certain discrete set time displacements. The timedomain version of the proposed P1-signal is

${{P\; 1_{j}(t)} = {K{\sum\limits_{k}{P\; 1_{j}(k){\mathbb{e}}^{2\;\pi\;{{\mathbb{i}}{({f + {{k \cdot \Delta}\; f}})}}t}}}}},$where for convenience we may include the frequency offset into f, andlet Δf be the spacing between two possible carriers of the P1-signal (=3times the subcarrier spacing of the 2 k OFDM-symbol). Assume that wehave a time error Δt that is less than the guard interval. Then the timedomain correlator sees

$\begin{matrix}{\left\langle {{P\; 1_{j}(t)},{P\; 1_{j}\left( {t + {\Delta\; t}} \right)}} \right\rangle = {K^{2}{\int_{period}^{\;}{\sum\limits_{k}^{\;}{\sum\limits_{k^{\prime}}\ {P\; 1_{j}(k)P\; 1_{j}\left( k^{\prime} \right){\mathbb{e}}^{2\;\pi\;{{\mathbb{i}}{({f + {{k \cdot \Delta}\; f}})}}t}}}}}}} \\{{\mathbb{e}}^{{- 2}\;\pi\;{{\mathbb{i}}{({f + {{k^{\prime} \cdot \Delta}\; f}})}}{({t + {\Delta\; t}})}}{\mathbb{d}t}} \\{= {K^{\prime}{\mathbb{e}}^{{- 2}\;\pi\;{\mathbb{i}}\; f\;\Delta\; t}{\sum\limits_{k}^{\;}{P\; 1_{j}(k)^{2}{\mathbb{e}}^{2\;\pi\;{\mathbb{i}}\;{k{({\Delta\;{f \cdot \;\Delta}\; t})}}}}}}}\end{matrix}$(hereafter referred to as equation (4)).

Here the coefficients K and K′ are there for normalization and containthe power boosting as well as the constants coming from DFT and theintegration. The absolute value of this term depends thus (up toscaling) only on the sum. Assume that Δt has such a magnitude that theproduct Δf Δt=n/511, for some integer n. That is, the time error is aninteger multiple of 1/511 of the common period of the subcarriers. So wecan write e^(2πjk(Δf·Δt))=ω^(nk). Taking into account the fact thatP1_(j)(k+1)=(1−e(αg^(k)))/2 only depends on the pattern of the comb (andnot at all on the BPSK-modulation) we see that at these values of thetime error the cross-correlation equals

$\left\langle {{P\; 1_{j}(t)},{P\; 1_{j}\left( {t + {n/\left( {{511 \cdot \Delta}\; f} \right)}} \right)}} \right\rangle = {K^{''}{\sum\limits_{k = 0}^{510}{\left( {1 - {{\mathbb{e}}\left( {\alpha\; g^{k}} \right)}} \right){\omega^{nk}.}}}}$(hereafter referred to as equation (5)).

The sums in equations (1) and (2) then tell us that (forgetting themultiplier K″—its absolute value is independent of n) this sum has thevalue 256, when n=0 (i.e. when there is no timing error) and hasabsolute value √{square root over (512)}≈22.6 otherwise. To summarize:with our signals there is a relatively dense discrete set of time errorsthat will lead to auto-correlations values about 10 dB below thesynchronized value. While this is not conclusive, it is highlysuggestive that the auto-correlation properties of our proposed signalsare relatively good.

Again the sums (1) and (2) are central to our estimate. When we comparetwo different P1-signals P1_(j) and P1_(j′), the computation that leadto equations (4) and (5) above will this time yield

$\left\langle {{P\; 1_{j}(t)},{P\; 1_{j^{\prime}}\left( {t + {n/\left( {{511 \cdot \Delta}\; f} \right)}} \right)}} \right\rangle = {K^{''}{\sum\limits_{k = 0}^{510}{\left( {1 - {{\mathbb{e}}\left( {\alpha\; g^{k}} \right)}} \right){{\mathbb{e}}\left( {\left( {\beta_{j} - \beta_{j^{\prime}}} \right)g^{k}} \right)}{\omega^{nk}.}}}}$

Recall that we work under the assumption that β_(j)−β_(j′)≠α. If heren=0, then this sum evaluates to 0 by formula (1), and otherwise we havehere two Gauss' sums, so by the triangle inequality we can estimate

${\left\langle {{P\; 1_{j}(t)},{P\; 1_{j^{\prime}}\left( {t + {n/\left( {{511 \cdot \Delta}\; f} \right)}} \right)}} \right\rangle } \leq {2K^{\prime\prime}{\sqrt{512}.}}$In other words, at this discrete set of time errors the crosscorrelations are at least 7 db below the perfect match of 256*K″.

Again the sum (2) allows us to give a relatively sharp estimate of theenvelope power at the sampling instants Δt=n/(511 Δf) for all n=0, 1, .. . , 510. We have

${{P\; 1_{j}\left( {n/\left( {{511 \cdot \Delta}\; f} \right)} \right)}} = {\frac{1}{\sqrt{2048}}{{{\sum\limits_{k = 0}^{510}{\left( {1 - {{\mathbb{e}}\left( {\alpha\; g^{k}} \right)}} \right){{\mathbb{e}}\left( {\beta_{j}g^{k}} \right)}\omega^{kn}}}}.}}$

As α≠β_(j) we get zero at n=0, and by the result of equation (2) onGauss' sums, the sum in absolute value signs is upper bounded by2√{square root over (512)}. Altogether the sampled envelope power isthus at most 1. Here the total signal energy is 256, so the mean poweris √{square root over (256/2048)}=1/√{square root over (8)}. Thus atthis (Nyquist) sampling rate the maximum to mean envelope power ratio isat most √{square root over (8)}. There is a general bound that tells usthat the continuous peak-to-mean-envelope-power-ratio is then at most (2ln(511)+1.132+4/511)√{square root over (8)} in the worst case (and inpractice most likely quite a bit better).

As discussed above, relatively fast recognition and synchronization ofOFDM signals may be achieved by using special synchronization signals orspecifically designed symbols, in accordance with various embodiments.For example, the P1 symbol may be defined to be a predetermined, e.g., 2k OFDM symbol, with a special structure using relatively sparselyallocated carriers (e.g., every third position is allowed, as discussedabove).

A potential issue with this approach is that, under certain multipathconditions, the energy of the synchronization symbol may besignificantly reduced, specifically on the selected active carriers. Forexample, if there is a two-path channel with 0 dB strength, and withdelay that is ⅓ of the OFDM symbol length (useful part without guardinterval), then every third carrier will be significantly cancelled.With a proper, unfortunate, phasing these cancellations may occur on theselected active carrier locations. The cancellation will be partial butcould still be potentially harmful.

Another potential issue is a continuous wave interference. This couldalso be difficult to handle with the approach set forth above.

Other approaches for synchronization symbols, like having a longersequence of known waveforms (like sinusoids) or pseudorandom codes, havebeen used in the past. The main drawback of such approaches is that theynormally use fairly long time periods and, hence, are relativelyinefficient. Data capacity is reduced. Also, a shorttraining/synchronization period is beneficial in mobile channels. Thenthe channel can remain roughly constant during the synchronizationsymbol, which increases the synchronization detection performance.

Embodiments use fairly short symbols (like 2 k or 1 k symbols in DVBcase), well defined in the frequency domain, including sparselydistributed carriers in pseudorandom locations, with relatively robustmodulation (like Binary Phase Shift Keying (BPSK) or Quadrature PhaseShift Keying (QPSK)). In addition, embodiments may use two consecutiveshort OFDM symbols, both carrying sparse active (“pilot”) carriers, withthe second symbol having the active carrier locations shifted by apredetermined amount in frequency. For implementation reasons, the shiftmay, in one embodiment, be one carrier interval of the OFDM symbol. Anembodiment takes advantage of the fact that only carrier locations whichare taken from a regular structure (every second, every third or thelike) are allowed. The active carriers have been distributed(pseudo)randomly on those locations. This leads to a regular structurein the time domain (known Fast Fourier Transform (FFT) samplingproperty): if every n^(th) carrier location is allowed, these positionsmay or may not have an active carrier, (other carriers being zero), thenthe OFDM symbol will have n identical consecutive parts in the timedomain.

FIG. 27 shows an example of a 2 k symbol (P1) symbol, in accordance withan embodiment, with length T_(u) 224 μs and with a guard interval oflength T_(g), which is ¼ of the useful symbol length T_(u). Ts, which isnot shown in FIG. 27 refers to the OFDM symbol length=Tu+Tg. Carrierlocations are taken from a raster of every third leading to periodicityof 3 in the useful signal part.

The periodicity shown in FIG. 27 can be used to detect the signalefficiently and reliably. The receiver may take a correlation betweencycle one and two to detect the signal existence. Note that in normaldata carrying OFDM symbols these correlations would be small—near zero.Detection robustness may be increased by taking multiple simultaneouscorrelations, e.g., take, in addition, correlation between cycle one andcycle three. It is possible to also add a correlation between cycle 2and cycle 3 for additional reliability.

Earlier solutions tend to rely on guard interval correlation (GIC) asthe data in the guard interval (cyclic prefix) is the same as the dataat the end of the useful symbol (end of cycle 3 in the example above).In accordance with embodiments, though, more signal energy is availablein correlation, and it is possible to use several simultaneouscorrelations over various parts of the signal that gives tolerancetowards interference and noise. In addition, a correlation between theguard interval and any of the n sections may also be performed.

In an additional aspect, some carrying of information may be includedinto the synchronization symbol. One possibility is to use a known BPSK(or QPSK) sequence in the OFDM symbols. The sequence may be one of a setof few sequences. These sequences may be the same or different for thefirst and second pulses. These sequences are correlated in the receiveragainst known sequences to determine which one was sent and, hence, afew bits of information may be carried. For instance, if the number ofBPSK possible sequence combinations is 8 then 3 bits may be carried.

A modification is that the sequence in symbol one may be used as areference. So the sequence in the first pilot symbol (P1a) is known tothe receiver in advance. The second pilot symbol could then have, say, mpossible sequences. The sequence in the symbol P1a is used as a roughchannel estimate for the symbol P1b. The receiver decides which sequencewas sent in P1b based on the phase information it got from the firstsymbol. Again, ld(m) (i.e., log₂(m)) bits are conveyed.

A further modification is that the information in sequence P1b may becoded using differential modulation so that the coded value in thesequence in P1b is given by the phase difference as compared to thephase value in the corresponding carrier position in the sequence inP1a. Typically this phase difference is 0 or 180 degrees in DifferentialBinary Phase Shift Keying (DBPSK). Differential Quadrature Phase ShiftKeying (DQPSK) may also be used.

In accordance with embodiments, a receiver may take advantage of theperiodicity that results from the sparse sub-sampling. The receiver maymake frequency translations in several (i.e., two or more) subsectionsof the received pulses, as is discussed in more detail below.

In accordance with embodiments, a transmitter may includesynchronization signals into the transmitted signal. One beneficialimplementation form is that the synchronization signal has a determinedform using a fixed size of FFT (e.g. 1 k in DVB-T2/H2). The datacarrying OFDM symbols following the synchronization symbols (there maybe some other synchronization signals also) could be of different formhaving different FFT size, symbol length, guard intervals (GI),modulation etc. This preferred structure of the synchronization symbolwill be described later. Second, the receiver could use the propertiesof the synchronization symbol in various ways. Some innovative receiveralgorithms will also be described.

The robustness of synchronization symbol P1 may be increased by dividingit into two parts, P1a and P1b, where both parts are 1 k FFT symbolswith relatively small guard intervals (GI) (like 1/16) or even withoutany GI. Short guard intervals are enough because, in any case, thedetection must be based on robustness of P1, and we cannot have longenough GIs to avoid Intersymbol interference (ISI). But boosting ofcarriers in P1 counteracts this. The boosting results from the fact thatonly a relatively low number of active carriers are used. In oneembodiment, a suitable value could be that, on average, every 6^(th)carrier is used. For 1 k FFT, this would mean on the order of 128pseudorandomly located active carriers. These active carriers may betaken from a raster where every nth carrier location is allowed, where nis a relatively small number, such as, 2, 3, 4, 5, or the like. The restof the carriers may have zero value.

FIG. 28 shows a synchronization symbol P1 having two consecutive OFDMsymbols (P1a, and P1b) that have the same FFT size in accordance with anembodiment. The active carrier locations are depicted by wide solidlines in FIG. 28. As shown in FIG. 28, the active carrier locations inP1b are shifted in frequency relative to the active carrier locations inP1a. Narrow solid horizontal lines in FIG. 28 represent allowed carrierlocations of the FFT symbol, and the dashed horizontal lines representforbidden carrier locations.

The location of the active carriers of the second part (P1b) would beshifted by some amount in frequency, in one embodiment, by one carrierlocation as compared to the first part P1a. This would give the benefitof not getting interference from P1a to P1b as the spill-over (due tomultipath propagation) from P1a would fall on the unused carrierlocations in P1b. Carrier locations 1-5 are labeled in FIG. 28. Carrierlocations 1, 3, 5, . . . are referred to as odd numbered carrierlocations and carrier locations 2, 4, 6, . . . are referred to as evennumbered carrier locations.

A few embodiments using examples for DVB-T2/H2 will now be provided. Ina first example, for robustness reasons, the structure of P1a could be:1 k FFT, GI length Tg= 1/16*Tu (corresponds to 64 samples). Basic rasterfor 128 carriers is ½, non-uniform pseudorandom pattern, active carrierswould have pseudorandom BPSK coding average ⅙ (e.g. only even carriersused). P1b could be otherwise similar but the locations shifted by one,i.e., only odd carrier locations used. The BPSK sequence could be thesame as, or different from, that in P1a. Various uncorrelated BPSKsequences may be used to indicate, e.g., the FFT size of the latercoming data symbols or some other useful parameters.

Additionally, in another aspect, the BPSK sequences in P1a and P1b couldbe selected so that the sequence in the first symbol P1a would remainthe same (and be known to the receiver). The extra information may becoded into the selection of the BPSK sequence of the second symbol P1b.The active carriers in the first symbol may then serve as pilot valuesfor the second symbol. As the symbol length is assumed to be shortrelative to the channel variations, the channel may be assumed to remainroughly unchanged for the time of the second symbol. Also, as thefrequency shift is one carrier interval (or some other small number),the channel phase may change by a relatively small amount from P1a toP1b (in the corresponding locations). So for BPSK signals, the phaseinformation would be close enough for fairly reliable detection (as thisis based on correlations over 128 carriers).

In a second DVB-T2/H2 example, as the sync symbol P1 is robust due tothe active carrier boosting (the average power may be the same as in theactual data symbols), guard intervals may be omitted. So it would bepossible to use the following structure: P1a: 1 k FFT, 0 GI, 128 activecarriers, pseudorandomly located, using only even carrier locations,BPSK modulated by a random sequence. P1b: same as P1a but active carrierlocations shifted up or down in frequency by one carrier frequencyinterval, i.e., using only odd carrier locations, if the P1a uses evencarrier locations. The FFT size information of the data symbols may becoded into the selection of the BPSK sequences. There may be, e.g., 5 or6 different sequences that should be maximally differing from eachother. The definition of these sequences could happen using normalcoding reasoning—maximizing the Hamming distance between the sequences.

In a third DVB-T2/H2 example, the pulse structure is the same as in thesecond example, but short guard intervals (like 1/16) are used in bothparts P1a and P1b.

Receiver algorithms in accordance with embodiments will now bediscussed. The pulse structure of P1, as described above, lends itselfto various detection algorithms. At least the following information maybe extracted from such a pulse structure: the existence of DVB-T2/H2 (orany other defined system) signal. As the signal has unique features inthe time domain (periodicity of n) and in the frequency domain (due tothe known BPSK sequence), the pilot symbol can be reliably detected witha relatively low probability of false detection.

Coarse timing of the signal may be obtained. The correlation propertiesof P1 give a good candidate for correct timing. The multiplecorrelations and/or utilization of most of the signal energy incorrelation make this reliable.

An estimate of the multipath channel delay profile may be obtained. Thetime correlation properties also give the rough structure of themultipath channel. Specifically for SFN networks, this could be used toestimate the difference of the delay extremes in the multipath profile(useful in locating the FFT window position for detecting the usefuldata).

An estimate of the frequency offset of the signal may be obtained. Thisis based on the FFT on the first symbol P1a and/or the second symbolP1b. Correlating, for example, the received powers on carriers with theknown active carrier locations, the offset could be solved.

Small amount of information coded into the selection of BPSK sequencesmay be conveyed (two methods as described above). For example, FFT sizeof the useful data symbols in the frame could be signalled.

For detection, half symbol correlation may be used and, preferably, forP1a and P1b simultaneously.

Carrier offset may be solved by taking 1 k FFT over P1a and/or P1b (P1bcould be used for extra robustness). Correlation with the knowncandidate BPSK patterns would solve FFT size of the useful data.

For SFN, a delay value of ½ of the 1 k symbol length could createpartial elimination of symbol P1a, for example. However, in that casethe symbol P1b would be unaffected (the delayed component addingconstructively). In principle, delay value of 1 symbol length couldcreate nulls for each 1 k carrier (for continuous sinusoids) but then itwould be harmless as the delayed interference would fall outside theuseful symbol period. So, consequently, delay values would not createnotable problems.

Aspects of an example receiver algorithm will now be discussed. For thespecial case of the second and third examples above, it is worth notingthat P1b is a transformed version of P1a. So if the receiver wouldtranslate pulse P1b by the known amount in frequency, then P1a and P1bwould be the same. That would mean that correlations could be taken overP1a and the frequency translated version of P1b (even with the guardinterval included). That might be beneficial to counteract someinterference like a CW interference on one frequency. The frequencytranslation would mean that such interference would practically bedivided into two parts that differ in frequency. The correlation of suchsignals would then be near zero and P1 could be detected reliablywithout significant influence from the interference.

Another aspect of an example possible receiver algorithm works in caseslike examples 2 and 3 above where the BPSK (or QPSK) sequences in P1aand P1b are the same.

FIG. 29 shows an example of P1 in which symbols P1a and P1b have eachbeen subdivided into two parts in accordance with an embodiment.Assuming that the allowed carrier positions are taken from a raster ofevery second (e.g., even carrier numbers for P1a and odd numbers forP1b) the useful part of the symbols include two identical parts (i.e.P1a1 and P1a2 and correspondingly P2a1 and P2a2). The receiver could nowtake correlations mixing parts of P1a and P1b (see FIG. 30). For thecorrelation purposes, the operations which span over the time intervalcovered by the total length of P1 the following arrangements may bemade: samples belonging to the first period including guard interval GI1and P1a1 (for the third example, this would mean 64+512=576 samples)will remain as they are; samples belonging to the second period (P1a2)will be frequency translated, i.e., each sample is multiplied by exp(j2π1/N) where N is the FFT size (1024 in this example) and i is theindex of the sample (576 . . . 1087), this will make the desiredfrequency shift by one carrier interval. Samples belonging to the firstpart of the second pulse including guard interval GI2 and P1b1 aretranslated in frequency to the opposite direction by multiplying thesesamples by exp(−j2π1/N), where i runs from 0 to 575. Samples belongingto P1b2 remain as they are.

The correlation is now formed by multiplying the modified samplescorresponding to the part P1a with the complex conjugate of the modifiedsamples corresponding to part P1b. The pair-wise multiplication resultsare summed together and the correlation result is achieved. The resultwill be maximized when the calculation period (in this example 2174samples long) falls on the received P1.

FIG. 30 is a schematic diagram of a correlator portion of a receiver inaccordance with an embodiment. The blocks labeled GI1-P1b2 represent thereceived P1 symbol of the OFDM signal frame. The data is fed into abuffer memory and pair-wise multiplications and summations take place ateach incoming sample time. The *'s in FIG. 30 represent complexconjugate. The expressions ‘Exp(jωt)’ and ‘Exp(−jωt)’ mean translationin frequency up and down, respectively.

FIG. 31 is a schematic diagram of a correlator portion of a receiver inaccordance with an embodiment. This arrangement leads to a narrowcorrelation peak. The pairwise correlation parts are different thanthose in the embodiment of FIG. 30. But the working principle andoperations are roughly the same. Guard intervals are not used incorrelation in the arrangement of FIG. 31. Again, the expressions‘Exp(jωt)’ and ‘Exp(−jωt)’ mean translation in frequency up and down,respectively.

In the arrangement of FIG. 31, the guard intervals are not used incorrelation, which makes the buffer memory slightly shorter.Significantly, the symbol parts that are pair-wise correlated are, onone hand, P1a1 vs. P1b2 and, on the other hand, P1a2 vs. P1b1. It can beshown that this will lead to a narrower correlation peak than thearrangement of FIG. 30. The drawback to the embodiment of FIG. 31 isthat the energy of guard intervals is not used. So it would bebeneficial to make GIs significantly shorter than the symbol length, oreven use zero GIs.

The embodiments of FIGS. 30 and 31 produce the following results: themodifications (translations) in P1a and P1b are symmetrical (frequencyshifts up and down), which might help in cancelling small identicalerrors in processing; the modifications take place at (about) half ofthe symbol length interval, which helps to “scramble” possiblecontinuous wave (CW), multipath, and other interference in correlation;whole pulse energy is utilized—also the guard interval in the embodimentof FIG. 30; and the correlation peak will be fairly narrow (only abouthalf of the width if P1a and P1b would be correlated in the way wherefrequency translation takes place in P1b (or P1a) only).

FIG. 32 shows steps of a detection sequence in accordance with anembodiment. FIG. 32 is presented in the context of processing a signalaccording to the third example set forth above.

A correlation is taken over a period that corresponds to the length ofsymbol P1a, as shown at 3202. The samples which are taken with delay ofTs (=length of P1a) are corrected by multiplying them with a complexcoefficient exp(−jωt) where ω is the frequency difference between theOFDM symbol carriers (=1/Tu).

The correlation result above is compared against a sliding average ofthe recent correlation results, as shown at 3204.

When the comparison result above exceeds a set threshold value, adecision is made that P1 is present, and the “yes” branch from 3206 isfollowed. Otherwise, processing for the correlation period is done, asshown at 3216.

The receiver determines a local maximum position of the correlation anddetermines the start of the P1 symbol based on the local maximumposition, as shown at 3208.

The receiver takes 1 k FFT over the P1a part of the signal, as shown at3210.

The receiver correlates various positions of BPSK sequences in thefrequency domain, as shown at 3212. The correlation maximum gives theposition of the frequency grid in the FFT domain that can used to solvethe difference between the receiver carrier frequency and the nominalfrequency (carrier offset). The sequence that maximizes the correlationindicates the FFT size of the actual data symbols.

Possible frequency offset is corrected for the detection of thefollowing symbols, as shown at 3214, and processing for the correlationperiod is then done, as shown at 3214.

In accordance with embodiments, tolerance against multipath fading isincreased as it would be highly improbable that both parts, P1a and P1bwould be cancelled at the same time. When detection in the receiver isbased on correlation results using both P1a and P1b, at least one resultwill be significantly higher than just correlating with noise or randomdata.

In addition, due to the periodicity of P1 caused by the sparse use ofcarrier positions, the receiver may base its decision regarding theexistence of a P1 signal on correlations over the periodic parts of thesignals. This periodicity increases the used signal power (aspractically whole signal energy can be used in detection) and alsoincreases the variety (diversity) as different parts of the signal intime domain can be used giving a variety of correlation results. Thesecorrelation results differ in interference (or unwanted signals) contentand can be used to make more reliable decisions.

One or more aspects of the invention may be embodied incomputer-executable instructions, such as in one or more programmodules, executed by one or more computers or other devices. Generally,program modules include routines, programs, objects, components, datastructures, etc. that perform particular tasks or implement particularabstract data types when executed by a processor in a computer or otherdevice. The computer executable instructions may be stored on a computerreadable medium such as a hard disk, optical disk, removable storagemedia, solid state memory, RAM, etc. As will be appreciated by one ofskill in the art, the functionality of the program modules may becombined or distributed as desired in various embodiments. In addition,the functionality may be embodied in whole or in part in firmware orhardware equivalents such as integrated circuits, field programmablegate arrays (FPGA), application specific integrated circuits (ASIC), andthe like.

Embodiments include any novel feature or combination of featuresdisclosed herein either explicitly or any generalization thereof. Whileembodiments have been described with respect to specific examplesincluding presently preferred modes of carrying out the invention, thoseskilled in the art will appreciate that there are numerous variationsand permutations of the above described systems and techniques. Thus,the spirit and scope of the invention should be construed broadly as setforth in the appended claims.

We claim:
 1. A method comprising: receiving a first orthogonal frequencydivision multiplexing pilot symbol; receiving a second orthogonalfrequency division multiplexing pilot symbol; frequency translating partof the first orthogonal frequency division multiplexing pilot symbol byone carrier interval in a first direction; frequency translating part ofthe second orthogonal frequency division multiplexing pilot symbol byone carrier interval in a second direction that is opposite from thefirst direction; and forming a correlation by multiplying the frequencytranslated part of the first orthogonal frequency division multiplexingpilot symbol by a complex conjugate of a part of the second orthogonalfrequency division multiplexing pilot symbol upon which frequencytranslation has not been performed, multiplying the frequency translatedpart of the second orthogonal frequency division multiplexing pilotsymbol by a complex conjugate of a part of the first orthogonalfrequency division multiplexing pilot symbol upon which frequencytranslation has not been performed, and summing the multiplicationresults.
 2. The method of claim 1, wherein frequency translating part ofthe first orthogonal frequency division multiplexing pilot symbol isperformed by multiplying each sample of the part being translated by exp(−j2πi/N), where N is a fast Fourier transform size of the firstorthogonal frequency division multiplexing pilot symbol and i is anindex of the sample.
 3. The method of claim 2, wherein frequencytranslating part of the first orthogonal frequency division multiplexingpilot symbol is performed by multiplying each sample of the guardinterval of the first orthogonal frequency division multiplexing pilotsymbol by exp (−j2πi/N).
 4. The method of claim 1, wherein frequencytranslating part of the second orthogonal frequency divisionmultiplexing pilot symbol is performed by multiplying each sample of thepart being translated by exp (j2πi/N), where N is a fast Fouriertransform size of the second orthogonal frequency division multiplexingpilot symbol and i is an index of the sample.
 5. The method of claim 4,wherein frequency translating part of the second orthogonal frequencydivision multiplexing pilot symbol is performed by multiplying eachsample of the guard interval of the second orthogonal frequency divisionmultiplexing pilot symbol by exp (j2πi/N).
 6. An apparatus comprising: aprocessor; and a memory storing executable instructions that, whenexecuted by the processor, cause the apparatus to: receive a firstorthogonal frequency division multiplexing pilot symbol, receive asecond orthogonal frequency division multiplexing pilot symbol,frequency translate part of the first orthogonal frequency divisionmultiplexing pilot symbol by one carrier interval in a first direction,frequency translate part of the second orthogonal frequency divisionmultiplexing pilot symbol by one carrier interval in a second directionthat is opposite from the first direction, and form a correlation bymultiplying the frequency translated part of the first orthogonalfrequency division multiplexing pilot symbol by a complex conjugate of apart of the second orthogonal frequency division multiplexing pilotsymbol upon which frequency translation has not been performed,multiplying the frequency translated part of the second orthogonalfrequency division multiplexing pilot symbol by a complex conjugate of apart of the first orthogonal frequency division multiplexing pilotsymbol upon which frequency translation has not been performed, andsumming the multiplication results.
 7. The apparatus of claim 6, whereinthe executable instructions are configured to, when executed by theprocessor, cause the apparatus to frequency translate part of the firstorthogonal frequency division multiplexing pilot symbol by multiplyingeach sample of the part being translated by exp (−j2πi/N), where N is afast Fourier transform size of the first orthogonal frequency divisionmultiplexing pilot symbol and i is an index of the sample.
 8. Theapparatus of claim 7, wherein the executable instructions are configuredto, when executed by the processor, cause the apparatus to frequencytranslate part of the first orthogonal frequency division multiplexingpilot symbol by multiplying each sample of the guard interval of thefirst orthogonal frequency division multiplexing pilot symbol by exp(−j2πi/N).
 9. The apparatus of claim 6, wherein the executableinstructions are configured to, when executed by the processor, causethe apparatus to frequency translate part of the second orthogonalfrequency division multiplexing pilot symbol by multiplying each sampleof the part being translated by exp (j2πi/N), where N is a fast Fouriertransform size of the second orthogonal frequency division multiplexingpilot symbol and i is an index of the sample.
 10. The apparatus of claim9, wherein the executable instructions are configured to, when executedby the processor, cause the apparatus to frequency translate part of thesecond orthogonal frequency division multiplexing pilot symbol bymultiplying each sample of the guard interval of the second orthogonalfrequency division multiplexing pilot symbol by exp (j2πi/N).
 11. Anon-transitory computer readable medium storing executable instructionsthat, when executed, cause a processor to perform operations comprising:receiving a first orthogonal frequency division multiplexing pilotsymbol; receiving a second orthogonal frequency division multiplexingpilot symbol; frequency translating part of the first orthogonalfrequency division multiplexing pilot symbol by one carrier interval ina first direction; frequency translating part of the second orthogonalfrequency division multiplexing pilot symbol by one carrier interval ina second direction that is opposite from the first direction; andforming a correlation by multiplying the frequency translated part ofthe first orthogonal frequency division multiplexing pilot symbol by acomplex conjugate of a part of the second orthogonal frequency divisionmultiplexing pilot symbol upon which frequency translation has not beenperformed, multiplying the frequency translated part of the secondorthogonal frequency division multiplexing pilot symbol by a complexconjugate of a part of the first orthogonal frequency divisionmultiplexing pilot symbol upon which frequency translation has not beenperformed, and summing the multiplication results.
 12. Thenon-transitory computer readable medium of claim 11, wherein frequencytranslating part of the first orthogonal frequency division multiplexingpilot symbol is performed by multiplying each sample of the part beingtranslated by exp (−j2πi/N), where N is a fast Fourier transform size ofthe first orthogonal frequency division multiplexing pilot symbol and iis an index of the sample.
 13. The non-transitory computer readablemedium of claim 12, wherein frequency translating part of the firstorthogonal frequency division multiplexing pilot symbol is performed bymultiplying each sample of the guard interval of the first orthogonalfrequency division multiplexing pilot symbol by exp (−j2πi/N).
 14. Thenon-transitory computer readable medium of claim 11, wherein frequencytranslating part of the second orthogonal frequency divisionmultiplexing pilot symbol is performed by multiplying each sample of thepart being translated by exp (j2πi/N), where N is a fast Fouriertransform size of the second orthogonal frequency division multiplexingpilot symbol and i is an index of the sample.
 15. The non-transitorycomputer readable medium of claim 14, wherein frequency translating partof the second orthogonal frequency division multiplexing pilot symbol isperformed by multiplying each sample of the guard interval of the secondorthogonal frequency division multiplexing pilot symbol by exp (j2πi/N).16. The method of claim 1, wherein at least one of the first orthogonalfrequency division multiplexing pilot symbol and the second orthogonalfrequency division multiplexing pilot symbol includes a subset ofavailable subcarriers configured for frequency offset estimation. 17.The method of claim 1, wherein the first orthogonal frequency divisionmultiplexing pilot symbol includes first active carriers distributed ina pseudorandom pattern; and the second orthogonal frequency divisionmultiplexing pilot symbol includes second active carriers in apseudorandom pattern.
 18. The method of claim 1, wherein at least one ofthe first orthogonal frequency division multiplexing pilot symbol andthe second orthogonal frequency division multiplexing pilot symbolincludes a pilot sequence constructed using at least one of thefollowing methods: pseudorandom sequencing, inverting according to apredetermined time, and boosting of a center carrier.
 19. The method ofclaim 1 wherein the first orthogonal frequency division multiplexingpilot symbol or the second orthogonal frequency division multiplexingpilot symbol includes a first carrier and a second carrier adjacent tothe first carrier, and information coded to a phase difference betweenthe first carrier and the second carrier.
 20. The method of claim 19,wherein the information coded to the phase difference results fromDifferential Binary Phase Shift Keying (DBPSK).